As they operate, digital circuits constantly switch the state of lines between a high-voltage

level and a low-voltage level to represent binary states. As shown in Figure 4.1*a*, the resulting

time-domain waveform on any single line of a digital circuit can thus be idealized as

a train of trapezoidal pulses of amplitude (either current *I* or voltage *V*) *A*, rise time *t*_{r}, fall

time *t*_{f}(between 10 and 90% of the amplitude), pulse width τ (at 50% of the amplitude),

and period *T*.

The Fourier envelope of all frequency-domain components generated by such a periodic

pulse train can be approximated by the nomogram of Figure 4.1*b*. The frequency spectrum

is composed mainly of a series of discrete sine-wave harmonics starting at the fundamental

frequency *f*_{0} = 1/*T *and continuing for all integer multiples of *f*_{0}. The nomogram

identifies two frequencies of interest. The first is *f*_{1}, above which the locus of the maximum

amplitudes rolls off with a 1/*f* slope. The second, *f*_{2}, is the limit above which the locus rolls

off at a more abrupt rate of 1/*f*^{2}. These frequencies are located at

| |

and

where *t* is the faster of (*t*_{r}, *t*_{f}).

The envelope of harmonic amplitude (in either amperes or volts) is then simplified to

For nonperiodic trains, the nomogram must be modified to account for the broadband

nature of the source. To do so, a nomogram of the spectral density envelope of the signal

can be defined for a unity bandwidth of 1 MHz by

Depending on its internal impedance, a circuit carrying such a pulse train will create in its

vicinity a field that is principally electric or magnetic. At a greater distance from the

source, the field becomes electromagnetic, regardless of the source impedance. If there is

a coupling mechanism, which can be either conduction or radiation, some or all of the

frequency components in the digital pulse train’s spectrum will be absorbed by some

“victim” receiver circuit.

To illustrate the magnitude of the problem, imagine a medical instrument’s main circuit

board, consisting of a CPU, some glue logic, and memory ICs, that has been housed in an

unshielded plastic case. Let’s assume that at any given time, a number of these ICs are toggling

states synchronously, at a frequency of 100 MHz, for instance. Furthermore, assume

that the total power switched at any given instant during a synchronous transition is

approximately 10 W. Now, in a real circuit, efficiency is not 100%, and a small fraction of

these 10 W will not do either useful work or be dissipated as heat by the ICs and wiring,

but rather, will be radiated into space. Assuming a reasonable fraction value of 10^{-6} of the

total switched power at the fundamental frequency, the power radiated is 10 μW.

Now, let’s assume that an FM radio is placed at a distance of 5 m from the device. The

field strength *E* produced by the 10 μW at this distance may be approximated by the

formula

Considering that the minimum field strength required for good reception quality by a typical

FM receiver is approximately 50 dBμV/m, the radiated computer clock would cause

considerable interference to the reception of a radio station in the same frequency. In fact,

interference caused by the computer of this example may extend up to 50 m or more away!

From the past discussion, it is easy to conclude that a first method for reducing radiated

emissions is to maintain clock speeds low as well as to make rise and fall times as slow as

possible for the specific application. At the same time, it is desirable to maintain the total

power per transition to the bare minimum. Transition times and powers depend primarily

on the technology used. As shown in Table 4.1, the ac parameters of each technology

strongly influence the equivalent radiation bandwidth. In addition, the voltage swing, in

combination with the source impedance and load characteristics of each technology, determines

the amount of power used and thus the power of radiated emissions on each transition.

Figure 4.2 shows how the selection of technology plays a crucial role in establishing

the bandwidth and power levels of radiated emissions that will require control throughout

the design effort.

Another problematic circuit often found in medical devices is the switching power supply.

Here, high-power switching at frequencies of 100 kHz and above produce significant

harmonics up to and above 30 MHz, requiring careful circuit layout and filtering. Fully

integrated filters are available for dc power lines. For example, Figure 4.3 shows the way

in which muRata BNX002 block filters are used to filter the raw dc power outputs produced

by two C&D Technologies’ HB04U15D12 isolating dc/dc converters. In the circuit,

each dc/dc (IC1 and IC2) produces unregulated 24 V (±12 V if the center-tap common is

used), which is isolated from the +15 V dc power input by an isolation barrier rated at

3000 V dc (continuous, tested at 8 kV, 60 Hz for 10 s). The outputs of the dc/dc converters

are filtered via filters FILT1 and FILT2, which internally incorporate multiple EMI filters

implemented with feed-through capacitors, monolithic chip capacitors, and ferrite-bead

inductors. Each of these filters attenuates RF by at least 40 dB in the range 1 MHz to

1 GHz. C1/C4 and C7/C10 are used to reduce ripple, and the circuits following these

capacitors are linear regulators that yield regulated ±24 V at 50 mA to the applied part for

which this isolation power supply was designed.

Another filter worth mentioning is muRata’s PLTxR53C common-mode choke coil.

This family of modules is ideal for suppressing noise from a few megahertz (1 to 5 MHz,

depending on the model) to several hundred megahertz (10 MHz to 1 GHz, depending on

the model) from dc power supplies. This module is useful in suppressing noise radiated

from the cable connecting a device to an external wall-mounted or “brick” ac adapter.

As they operate, digital circuits constantly switch the state of lines between a high-voltage

level and a low-voltage level to represent binary states. As shown in Figure 4.1*a*, the resulting

time-domain waveform on any single line of a digital circuit can thus be idealized as

a train of trapezoidal pulses of amplitude (either current *I* or voltage *V*) *A*, rise time *t*_{r}, fall

time *t*_{f}(between 10 and 90% of the amplitude), pulse width τ (at 50% of the amplitude),

and period *T*.

The Fourier envelope of all frequency-domain components generated by such a periodic

pulse train can be approximated by the nomogram of Figure 4.1*b*. The frequency spectrum

is composed mainly of a series of discrete sine-wave harmonics starting at the fundamental

frequency *f*_{0} = 1/*T *and continuing for all integer multiples of *f*_{0}. The nomogram

identifies two frequencies of interest. The first is *f*_{1}, above which the locus of the maximum

amplitudes rolls off with a 1/*f* slope. The second, *f*_{2}, is the limit above which the locus rolls

off at a more abrupt rate of 1/*f*^{2...}

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